The present invention relates to a switching power supply circuit to be provided as a power supply for various electronic apparatus.
Switching power supply circuits employing switching converters such as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Hence, various switching power supply circuits using various resonance type converters have been previously proposed by the present applicant. A resonance type converter makes it possible to readily obtain high power conversion efficiency, and to achieve low noise because the resonance type converter forms a sine-wave waveform in switching operation. The resonance type converter has another advantage of being able to be formed by a relatively small number of parts.
FIG. 9 is a circuit diagram showing an example of a prior art switching power supply circuit that can be formed according to an invention previously proposed by the present applicant.
The power supply circuit shown in FIG. 9 is provided with a full-wave rectifier circuit comprising a bridge rectifier circuit Di and a smoothing capacitor Ci. The full-wave rectifier circuit serves as a rectifying and smoothing circuit supplied with a commercial alternating-current power (alternating-current input voltage VAC) to provide a direct-current input voltage. The rectifying and smoothing circuit generates a rectified and smoothed voltage Ei whose level is equal to that of the alternating-current input voltage VAC multiplied by unity.
A voltage resonance type converter that includes a switching device Q1 and performs switching operation by a so-called single-ended system is provided as a switching converter for interrupting the rectified and smoothed voltage Ei (direct-current input voltage) inputted from the rectifying and smoothing circuit.
The voltage resonance type converter in this case is externally excited, and a MOS-FET, for example, is used as the switching device Q1. A drain of the switching device Q1 is connected to a positive electrode of a smoothing capacitor Ci via a primary winding N1 of an insulating converter transformer PIT, while a source of the switching device Q1 is connected to a ground on the primary side.
A parallel resonant capacitor Cr is connected in parallel with the drain and source of the switching device Q1. Capacitance of the parallel resonant capacitor Cr and leakage inductance obtained at the primary winding N1 of the insulating converter transformer PIT form a primary-side parallel resonant circuit. The parallel resonant circuit performs resonant operation according to switching operation of the switching device Q1. Thus, the switching operation of the switching device Q1 is of a voltage resonance type.
Also, a clamp diode (so-called body diode) DD is connected in parallel with the drain and source of the switching device Q1. The clamp diode DD forms a path of clamp current that flows during an off period of the switching device.
In this case, the drain of the switching device Q1 is connected to an oscillating circuit 41 in a switching driver 10B, which will be described next. An output of the drain supplied to the oscillating circuit 41 is used in switching frequency control to variably control an on period within one switching cycle, which will be described later.
The switching device Q1 is driven for switching operation by the switching driver 10B which is formed by integrating the oscillating circuit 41 and a driving circuit 42, and the switching frequency of the switching device Q1 is variably controlled for the purpose of constant-voltage control. Incidentally, the switching driver 10B in this case is provided as a single integrated circuit (IC), for example.
The switching d river 10B is connected to a line of the rectified and smoothed voltage Ei via a starting resistance RS. The switching driver 10B starts operation by being supplied with the power supply voltage via the starting resistance Rs at the start of power supply, for example.
The oscillating circuit 41 in the switching driver 10B performs oscillating operation to generate and output an oscillating signal. The driving circuit 42 converts the oscillating signal into a driving voltage, and then outputs the driving voltage to a gate of the switching device Q1. Thus, the switching device Q1 performs switching operation according to the oscillating signal generated by the oscillating circuit 41. Therefore, the switching frequency and duty ratio of an on/off period within one switching cycle of the switching device Q1 is determined depending on the oscillating signal generated by the oscillating circuit 41.
The oscillating circuit 41 changes the frequency of the oscillating signal (switching frequency fs) on the basis of the level of a secondary-side direct-current output voltage E0 inputted via a photocoupler 40, which will be described later. The oscillating circuit 41 changes the switching frequency fs and at the same time, controls the waveform of the oscillating signal in such a manner that a period TOFF during which the switching device Q1 is turned off is fixed and a period TON (conduction angle) during which the switching device Q1 is turned on is changed. The period TON (conduction angle) is variably controlled on the basis of the peak value of a parallel resonance voltage V1 across the parallel resonant capacitor Cr. As a result of such operation of the oscillating circuit 41, the secondary-side direct-current output voltage E0 is stabilized, as will be described later.
The insulating converter transformer PIT transmits switching output of the switching device Q1 to the secondary side of the switching power supply circuit.
As shown in FIG. 11, the insulating converter transformer PIT has an E-E-shaped core formed by combining E-shaped cores CR1 and CR2 made for example of a ferrite material in such a manner that magnetic legs of the core CR1 are opposed to magnetic legs of the core CR2. A primary winding N1 and a secondary winding N2 are wound around a central magnetic leg of the E-E-shaped core in a state divided from each other by using a dividing bobbin B. Also, a gap G is formed in the central magnetic leg, as shown in FIG. 11, to provide loose coupling at a required coupling coefficient.
The gap G can be formed by making the central magnetic leg of each of the E-shaped cores CR1 and CR2 shorter than two outer legs of each of the E-shaped cores CR1 and CR2. The coupling coefficient k is set for example to be k≈0.85 to provide a loosely coupled state, whereby a saturated state is not readily obtained.
As shown in FIG. 9, an ending point of the primary winding N1 of the insulating converter transformer PIT is connected to the drain of the switching device Q1, while a starting point of the primary winding N1 is connected to the positive electrode of the smoothing capacitor Ci (rectified and smoothed voltage Ei). Hence, the primary winding N1 is supplied with the switching output of the switching device Q1, whereby an alternating voltage whose cycle corresponds to the switching frequency of the switching device Q1 occurs in the primary winding N1.
An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 on the secondary side of the insulating converter transformer PIT. In this case, a secondary-side parallel resonant capacitor C2 is connected in parallel with the secondary winding N2, and therefore leakage inductance L2 of the secondary winding N2 and capacitance of the secondary-side parallel resonant capacitor C2 form a parallel resonant circuit. The parallel resonant circuit converts the alternating voltage induced in the secondary winding N2 into a resonance voltage, whereby voltage resonance operation is obtained on the secondary side.
Thus, the power supply circuit is provided with the parallel resonant circuit to convert switching operation into voltage resonance type operation on the primary side, and the parallel resonant circuit to provide voltage resonance operation on the secondary side. In the present specification, the switching converter provided with resonant circuits on the primary side and the secondary side is also referred to as a xe2x80x9ccomplex resonance type switching converter.xe2x80x9d
A rectifying and smoothing circuit comprising a bridge rectifier circuit DBR and a smoothing capacitor C0 is provided on the secondary side of the power supply circuit formed as described above, whereby a secondary-side direct-current output voltage E0 is obtained. This means that according to the configuration of the power supply circuit, full-wave rectifying operation on the secondary side is provided by the bridge rectifier circuit DBR. In this case, the bridge rectifier circuit DBR is supplied with the resonance voltage by the secondary-side parallel resonant circuit, and then generates the secondary-side direct-current output voltage E0 whose level is substantially equal to that of the alternating voltage induced in the secondary winding N2.
The secondary-side direct-current output voltage E0 is also inputted to the oscillating circuit 41 in the switching driver 10B on the primary side via the photocoupler 40 insulating the primary side from the secondary side.
As for secondary-side operation of the insulating converter transformer PIT, mutual inductance M between inductance L1 of the primary winding N1 and inductance L2 of the secondary winding N2 becomes +M or xe2x88x92M, depending on polarity (winding direction) of the primary winding N1 and the secondary winding N2, a connecting relation of a rectifier diode D0 (D01, D02), and change in polarity of the alternating voltage induced in the secondary winding N2.
For example, an equivalent of a circuit shown in FIG. 12A has a mutual inductance of +M, while an equivalent of a circuit shown in FIG. 12B has a mutual inductance of xe2x88x92M.
This will be applied to the secondary-side operation of the insulating converter transformer PIT shown in FIG. 9; when the alternating voltage obtained at the secondary winding N2 has a positive polarity, an operation that causes rectified current to flow in the bridge rectifier circuit DBR can be considered a +M operation mode (forward operation), whereas when the alternating voltage obtained at the secondary winding N2 has a negative polarity, an operation that causes rectified current to flow in the bridge rectifier diode DBR can be considered a xe2x88x92M operation mode (flyback operation). As the alternating voltage obtained at the secondary winding N2 changes from positive polarity to negative polarity and vice versa, the operation mode of the mutual inductance changes from +M to xe2x88x92M and vice versa, respectively.
With such a configuration, power increased by effects of the primary-side parallel resonant circuit and the secondary-side parallel resonant circuit is supplied to a load side, and accordingly the power supplied to the load side is increased as much, thereby improving a rate of increase of maximum load power.
This is achieved because as described earlier with reference to FIG. 11, the gap G is formed in the insulating converter transformer PIT to provide loose coupling at a required coupling coefficient, and thereby a saturated state is not readily obtained. For example, when the gap G is not provided in the insulating converter transformer PIT, it is highly likely that the insulating converter transformer PIT reaches a saturation state and then performs abnormal flyback operation. Therefore it is difficult to expect that the full-wave rectifying operation mentioned above will be properly performed.
Stabilizing operation of the circuit shown in FIG. 9 is as follows.
As described above, the oscillating circuit 41 in the switching driver 10B on the primary side is supplied with the secondary-side direct-current output voltage E0 via the photocoupler 40. The oscillating circuit 41 changes the frequency of the oscillating signal for output according to change in the level of the supplied secondary-side direct-current output voltage E0. This means an operation of changing the switching frequency of the switching device Q1. Thus, resonance impedance of the primary-side voltage resonance type converter and the insulating converter transformer PIT is changed, and accordingly energy transmitted to the secondary side of the insulating converter transformer PIT is also changed. As a result, the secondary-side direct-current output voltage E0 is controlled so as to remain constant at a required level. This means that the power supply is stabilized.
When the oscillating circuit 41 of the power supply circuit shown in FIG. 9 changes the switching frequency, the period TOFF during which the switching device Q1 is turned off is fixed and the period TON during which the switching device Q1 is turned on is variably controlled, as described above. Specifically, by variably controlling the switching frequency as an operation for constant-voltage control, the power supply circuit controls the resonance impedance for switching output, and at the same time, controls the conduction angle of the switching device within a switching cycle (PWM control). This complex control operation is realized by a single control circuit system. In the present specification, such complex control is also referred to as xe2x80x9ca complex control method.xe2x80x9d
FIG. 10 is a circuit diagram showing another example of a power supply circuit formed according to an invention previously proposed by the present applicant. In the figure, the same parts as in FIG. 9 are identified by the same reference numerals, and their description will be omitted.
A self-excited voltage resonance type converter circuit that performs single-ended operation by a switching device Q1 is provided on the primary side of the power supply circuit of FIG. 10. In this case, a high voltage bipolar transistor (BJT; junction transistor) is employed as the switching device Q1.
A base of the switching device Q1 is connected to a positive electrode side of a smoothing capacitor Ci (rectified and smoothed voltage Ei) via a base current limiting resistance RB and a starting resistance RS, so that base current at the start of power supply is taken from a line of a rectifying and smoothing circuit. Connected between the base of the switching device Q1 and a primary-side ground is a series resonant circuit for self-oscillation driving that is formed by connecting a driving winding NB, a resonant capacitor CB, and the base current limiting resistance RB in series with each other.
A clamp diode DD inserted between the base of the switching device Q1 and a negative electrode of the smoothing capacitor Ci (primary-side ground) forms a path of clamp current that flows during the off period of the switching device Q1. A collector of the switching device Q1 is connected to one end of a primary winding N1 of an insulating converter transformer PIT, while an emitter of the switching device Q1 is grounded.
A parallel resonant capacitor Cr is connected in parallel with the collector and emitter of the switching device Q1. Also in this case, capacitance of the parallel resonant capacitor Cr and leakage inductance L1 of the primary winding N1 side of the insulating converter transformer PIT form a primary-side parallel resonant circuit of the voltage resonance type converter.
An orthogonal type control transformer PRT shown in FIG. 10 is a saturable reactor provided with a resonance current detecting winding ND, the driving winding NB, and a control winding NC. The orthogonal type control transformer PRT is provided to drive the switching device Q1 and effect control for constant voltage.
The structure of the orthogonal type control transformer PRT is a cubic core, not shown in the figure, formed by connecting two double-U-shaped cores each having four magnetic legs with each other at ends of the magnetic legs. The resonance current detecting winding ND and the driving winding NB are wound around two given magnetic legs of the cubic core in the same winding direction, and the control winding NC is wound in a direction orthogonal to the resonance current detecting winding ND and the driving winding NB.
In this case, the resonance current detecting winding ND of the orthogonal type control transformer PRT is inserted in series between the positive electrode of the smoothing capacitor Ci and the primary winding N1 of the insulating converter transformer PIT, so that the switching output of the switching device Q1 is transmitted to the resonance current detecting winding ND via the primary winding N1. The switching output obtained by the resonance current detecting winding ND of the orthogonal type control transformer PRT is induced in the driving winding NB via transformer coupling, whereby an alternating voltage is generated as driving voltage in the driving winding NB. The driving voltage is outputted as driving current to the base of the switching device Q1 from a series resonant circuit (NB and CB), which forms the self-oscillation driving circuit, via the base current limiting resistance RB. Thus, the switching device Q1 performs switching operation at a switching frequency determined by the resonance frequency of the series resonant circuit. Then the switching output obtained at the collector of the switching device Q1 is transmitted to the primary winding N1 of the insulating converter transformer PIT.
The insulating converter transformer PIT provided in the circuit of FIG. 10 has the same structure as described earlier with reference to FIG. 11, thus providing loose coupling between the primary side and the secondary side.
A secondary-side parallel resonant capacitor C2 is connected in parallel with a secondary winding N2 on the secondary side of the insulating converter transformer PIT in the circuit of FIG. 10 to thereby form a secondary-side parallel resonant circuit. Thus, the power supply circuit also has a configuration of a complex resonance type switching converter.
A half-wave rectifier circuit comprising a diode D0 and a smoothing capacitor C0 is provided for the secondary winding N2 on the secondary side of the power supply circuit, so that a secondary-side direct-current output voltage E0 is obtained by half-wave rectifying operation that comprises only a forward operation. In this case, the secondary-side direct-current output voltage E0 is also inputted from a branch point to a control circuit 1, and the control circuit 1 uses the direct-current output voltage E0 as a detection voltage.
The control circuit 1 variably controls inductance LB of the driving winding NB wound in the orthogonal type control transformer PRT by changing the level of a control current (direct current) flowing through the control winding NC according to change in the level of the secondary-side direct-current output voltage E0. This results in a change in resonance conditions of the series resonant circuit including the inductance LB of the driving winding NB in the circuit for self-oscillation driving of the switching device Q1. This means an operation of changing the switching frequency of the switching device Q1, by which the secondary-side direct-current output voltage is stabilized. Also in such a configuration for constant-voltage control including the orthogonal type control transformer PRT, the switching converter on the primary side is of voltage resonance type, and therefore the power supply circuit performs operation by the complex control method, in which the power supply circuit variably controls the switching frequency and at the same time controls the conduction angle of the switching device within a switching cycle (PWM control).
FIGS. 13A to 13F are waveform diagrams showing operation of the power supply circuit shown in FIG. 10. FIGS. 13A, 13B, and 13C each show operation of the power supply circuit at an alternating-current input voltage VAC=100 V and a maximum load power Pomax=200 W. FIGS. 13D, 13E, and 13F each show operation of the power supply circuit at an alternating-current input voltage VAC=100 V and a minimum load power Pomin=0 W, or no load.
When the switching device Q1 performs switching operation on the primary side, the primary-side parallel resonant circuit performs resonant operation during the period TOFF during which the switching device Q1 is turned off. Thus, as shown in FIGS. 13A and 13D, a parallel resonance voltage V1 across the parallel resonant capacitor Cr forms a sinusoidal resonance pulse waveform during the period TOFF. In the case of the complex resonance type converter having a parallel resonant circuit as a secondary-side resonant circuit, the period TOFF during which the switching device Q1 is turned off is fixed, while the period TON during which the switching device Q1 is turned on is changed, as shown in the figures.
The voltage resonance type converter on the primary side performs switching operation at the timing described above, and thereby the rectifier diode D0 on the secondary side performs switching and rectifying operation on the alternating voltage induced in the secondary winding N2.
In this case, as shown in FIGS. 13B and 13E, a voltage Vo across the secondary winding N2 is clamped at a level of the secondary-side direct-current output voltage E0 during a period DON during which the rectifier diode D0 is turned on, while the voltage Vo forms a sinusoidal pulse waveform in a direction of negative polarity due to resonance effect of the secondary-side parallel resonant circuit during a period DOFF during which the rectifier diode D0 is turned off. As shown in FIGS. 13C and 13F, a secondary-side rectified current I0 to be stored in the smoothing capacitor C0 via the rectifier diode D0 steeply rises at the start of the period DON and thereafter gradually lowers its level, thus forming substantially a sawtooth waveform.
A comparison of FIG. 13A with FIG. 13D indicates that switching frequency fs is controlled so as to rise as load power Po is decreased, and the switching frequency fs (switching cycle) is changed while fixing the period TOFF at a constant length and changing the period TON, during which the switching device Q1 is turned on. This represents operation by the above-mentioned complex control method.
The voltage resonance type converter formed as shown in FIG. 10 changes the level of the parallel resonance voltage V1 according to variation in load power. For example, the parallel resonance voltage V1 is 550 Vp at a maximum load power Pomax=200 W, whereas the parallel resonance voltage V1 becomes 300 Vp at a minimum load power Pomin=0 W. This means that the parallel resonance voltage V1 has a tendency to rise as the load power becomes heavier.
Similarly, the peak level of the voltage Vo across the secondary winding N2 obtained during the period DOFF has a tendency to rise as the load power becomes heavier. In this case, the voltage Vo is 450 Vp at a maximum load power Pomax=200 W, whereas the voltage Vo is 220 Vp at a minimum load power Pomin=0 W.
Incidentally, the circuit shown in FIG. 9 performs substantially the same operation as described by using the waveform diagrams of FIGS. 13A to 13F.
Next, as characteristics of the power supply circuits shown in FIGS. 9 and 10, FIG. 14 shows characteristics of variations in the switching frequency fs, the period TOFF and the period TON within a switching cycle, and the parallel resonance voltage V1 with respect to the alternating-current input voltage VAC at a maximum load power Pomax=200 W.
First, FIG. 14 shows that the switching frequency fs is changed within a range of fs=110 KHz to 140 KHz for the alternating-current input voltage VAC=90 V to 140 V. This indicates an operation of stabilizing variation in the secondary-side direct-current output voltage E0 according to variation in direct-current input voltage. As for variation in the alternating-current input voltage VAC, the switching frequency is controlled so as to rise as the level of the alternating-current input voltage VAC is increased.
As for the period TOFF and the period TON within one switching cycle, the period TOFF is constant, as contrasted with the switching frequency fs, whereas the period TON is reduced so as to form a quadratic curve as the switching frequency fs is increased. This also indicates an operation for controlling the switching frequency by the complex control method.
The parallel resonance voltage V1 also changes according to variation in commercial alternating-current power VAC; as shown in FIG. 14, the level of the parallel resonance voltage V1 rises as the alternating-current input voltage VAC is increased.
FIG. 15 shows another power supply circuit according to an invention previously proposed by the present applicant. The power supply circuit shown in FIG. 15 is a complex resonance type switching converter provided with a series resonant circuit on the secondary side. In the figure, the same parts as in FIGS. 9 and 10 are identified by the same reference numerals, and their description will be omitted.
The power supply circuit of FIG. 15 has the following configuration on the secondary side.
A starting point of a secondary winding N2 of an insulating converter transformer PIT is connected to a node of an anode of a rectifier diode D01 and a cathode of a rectifier diode D02 via a series resonant capacitor Cs connected in series with the secondary winding N2, while an ending point of the secondary winding N2 is connected to a ground on the secondary side. A cathode of the rectifier diode D01 is connected to a positive electrode of a smoothing capacitor C0, and an anode of the rectifier diode D02 is connected to the secondary-side ground. A negative electrode of the smoothing capacitor C0 is connected to the secondary-side ground.
Such a connection forms a voltage doubler half-wave rectifier circuit comprising a set of the [secondary winding N2, series resonant capacitor Cs, rectifier diodes D01 and D02, and smoothing capacitor C0].
Capacitance of the series resonant capacitor Cs and leakage inductance L2 of the secondary winding N2 form a series resonant circuit that performs resonant operation in response to on/off operation of the rectifier diodes D01 and D02.
The capacitance of the series resonant capacitor Cs is selected so as to satisfy fo1≈fo2, where parallel resonance frequency of a parallel resonant circuit (N1 and Cr) on the primary side is fo1, and series resonance frequency of the series resonant circuit on the secondary side is fo2.
Thus, the power supply circuit of FIG. 15 is a xe2x80x9ccomplex resonance type switching converterxe2x80x9d provided with the parallel resonant circuit to convert switching operation into voltage resonance type operation on the primary side, and the series resonant circuit to provide current resonance operation on the secondary side.
Operation of the voltage doubler rectifier circuit comprising a set of the [secondary winding N2, series resonant capacitor Cs, rectifier diodes D01 and D02, and smoothing capacitor C0] is as follows, for example.
When switching output is obtained at a primary winding N1 as a result of switching operation on the primary side, the switching output is induced at the secondary winding N2. The voltage doubler rectifier circuit is supplied with an alternating voltage obtained in the secondary winding N2 to perform rectifying operation thereon.
In this case, during a period during which the rectifier diode D01 is turned off and the rectifier diode D02 is turned on, the voltage doubler rectifier circuit performs operation in subtractive polarity mode, in which polarity of the primary winding N1 and the secondary winding N2 is xe2x88x92M, and thereby stores a current rectified by the rectifier diode D02 in the series resonant capacitor Cs.
During a period during which the rectifier diode D02 is turned off and the rectifier diode D01 is turned on, the voltage doubler rectifier circuit performs operation in additive polarity mode, in which polarity of the primary winding N1 and the secondary winding N2 is +M, and thereby stores in the smoothing capacitor C0 a current obtained by adding potential of the series resonant capacitor Cs to the voltage induced in the secondary winding N2.
The rectifying operations in subtractive polarity mode and additive polarity mode described above are alternately repeated as the secondary-side series resonant circuit performs series resonant operation on the secondary side of the insulating converter transformer PIT. As a result, the smoothing capacitor C0 obtains a secondary-side direct-current output voltage E0 that has a level substantially twice that of the voltage induced in the secondary winding N2.
Since in this case, the secondary-side direct-current output voltage E0 is obtained by the operation of the voltage doubler rectifier circuit, the number of turns of the secondary winding N2 needs to be only about xc2xd of that in a configuration having for example an equal-voltage rectifier circuit on the secondary side.
Also in this case, the secondary-side direct-current output voltage E0 is fed back to an oscillating circuit 41 in a switching driver 10B on the primary side via a photocoupler 40, and on the basis of the fed-back secondary-side direct-current output voltage E0, constant-voltage operation by the complex control method is obtained on the primary side.
Next, FIG. 16 shows another example of a complex resonance type switching converter provided with a series resonant circuit on the secondary side. As in the case of the power supply circuit shown in FIG. 10, the power supply circuit of FIG. 16 is provided with a self-excited voltage resonance type converter that performs single-ended operation on the primary side.
Also in this case, a series resonant capacitor Cs is connected in series with a starting point of a secondary winding N2 on the secondary side of the power supply circuit to form a secondary-side series resonant circuit. The power supply circuit in this case is provided with a bridge rectifier circuit DBR as a secondary-side rectifier circuit. The starting point of the secondary winding N2 is connected to a positive electrode input terminal of the bridge rectifier circuit DBR via the series resonant capacitor Cs, and a ending point of the secondary winding N2 is connected to a negative electrode input terminal of the bridge rectifier circuit DBR. In this circuit configuration, an alternating voltage obtained in the secondary winding N2, that is, resonance output of the secondary-side series resonant circuit is subjected to full-wave rectification by the bridge rectifier circuit DBR, and then stored in a smoothing capacitor C0, whereby a secondary-side direct-current output voltage E0 is obtained.
Also in this case, the secondary-side direct-current output voltage E0 is inputted from a branch point to a control circuit 1, and the control circuit 1 uses the inputted direct-current output voltage E0 as a detection voltage for constant-voltage control.
FIGS. 17A to 17F are waveform diagrams showing operation of the power supply circuits shown in FIGS. 15 and 16. FIGS. 17A, 17B, and 17C each show operation of the power supply circuits at an alternating-current input voltage VAC=100 V and a maximum load power Pomax=200 W. FIGS. 17D, 17E, and 17F each show operation of the power supply circuits at an alternating-current input voltage VAC=100 V and a minimum load power Pomin=0 W, or no load.
As shown in FIGS. 17A and 17D, a parallel resonance voltage V1 obtained across the parallel resonant capacitor Cr by switching operation of a switching device Q1 forms a sinusoidal resonance pulse waveform during a period TOFF. In the case of the complex resonance type converter having a parallel resonant circuit as a secondary-side resonant circuit, the period TOFF during which the switching device Q1 is turned off is changed, as shown in the figures.
Waveforms of FIGS. 17A and 17D show that also in this case, switching frequency fs is controlled so as to rise as load power Po is decreased. Also, the switching frequency fs (switching cycle) is changed by varying a period TON during which the switching device Q1 is turned on within one switching cycle.
The circuits formed as shown in FIGS. 15 and 16 have a tendency to raise the level of the parallel resonance voltage V1 as the load power becomes heavier. In this case, the parallel resonance voltage V1 is 580 Vp at a maximum load power Pomax=200 W, whereas the parallel resonance voltage V1 is 380 Vp at a minimum load power Pomin=0 W.
As shown in FIGS. 17B and 17E, a switching output current IQ1 flowing through a drain or a collector of the switching device Q1 is in synchronism with timing of the periods TOFF and TON, and forms substantially the same waveform pattern as that shown in FIGS. 13B and 13E. Specifically, the switching output current IQ1 is at a zero level during the period TOFF, and the switching output current IQ1 flows in a manner shown by the waveforms of FIGS. 17B and 17E during the period TON. Also in the case of this circuit configuration, the switching output current IQ1 has a tendency to increase as the load power Po becomes heavier. In this case, the switching output current IQ1 is 3.6 A at a maximum load power Pomax=200 W, whereas the switching output current IQ1 is 0.3 A at a minimum load power Pomin=0 W.
Operation on the secondary side is shown as a voltage V1 across the secondary winding N2 in FIGS. 13C and 13F. According to the figures, the voltage provides a rectangular pulse clamped at the level of the secondary-side direct-current output voltage E0 during the period DON at a maximum load power Pomax=200 W, while at a minimum load power Pomin=0 W, the voltage provides a sine wave having a switching cycle of the primary side, whose peak level is clamped at the level of the secondary-side direct-current output voltage E0.
As characteristics of the power supply circuits shown in FIGS. 15 and 16, FIG. 18 shows characteristics of variations in the switching frequency fs, the period TOFF and the period TON within a switching cycle, and the parallel resonance voltage V1 with respect to alternating-current input voltage VAC at a maximum load power Pomax=200 W.
FIG. 18 shows that the switching frequency fs is changed within a range of fs=110 KHz to 160 KHz for the alternating-current input voltage VAC=90 V to 140 V. This indicates an operation of stabilizing variation in the secondary-side direct-current output voltage E0 according to variation in direct-current input voltage. Also in this case, the switching frequency is controlled so as to rise as the level of the alternating-current input voltage VAC is increased.
Under conditions of a constant load, for example, the period TOFF within one switching cycle is constant, as contrasted with the switching frequency fs, whereas the period TON within one switching cycle is reduced as the switching frequency fs is increased. This also indicates an operation for controlling the switching frequency by the complex control method.
As shown in FIG. 18, in this case, the parallel resonance voltage V1 changed according to variation in commercial alternating-current power VAC is lowered within a level range around 600 V as the alternating-current input voltage VAC is increased within a range of the alternating-current input voltage VAC=80 to 100 V, and the parallel resonance voltage V1 is raised for the alternating-current input voltage VAC=100 V or higher.
The power supply circuit s shown in FIGS. 9, 10, 15, and 16 have the following problems.
The externally excited voltage resonance type converter employing the complex control method as a constant-voltage control method, as shown in FIGS. 9 and 15, uses a variation component in the level of the secondary-side direct-current output voltage E0 to control the switching frequency on the primary side, and therefore, in practice, the level of the secondary-side direct-current output voltage E0 is detected and amplified on the secondary side and then the result is supplied to the switching driver on the primary side, for example. Also, in this case, it is necessary to insulate the primary side from the secondary side by providing a photocoupler or the like between the primary side and the secondary side. In addition, the peak value of the parallel resonance voltage on the primary side is detected for conduction angle control of the primary-side voltage resonance type converter. Such a configuration renders an actual power supply circuit complex and large.
In the case of the self-excited voltage resonance type converter employing the complex control method as shown in FIGS. 10 and 16, it is necessary to provide insulation distance between the current detecting winding ND, the driving winding NB, and the cores provided in the orthogonal type control transformer PRT, which is required for constant-voltage control. This makes it difficult to design and manufacture the orthogonal type control transformer PRT and also hinders miniaturization of the orthogonal type control transformer PRT.
Since in the circuits of FIGS. 9 and 10 provided with a secondary-side parallel resonant circuit, the voltage Vo across the secondary winding N2 rises to about 450 Vp at the maximum, components having a withstand voltage of about 600 V, for example, are to be selected as the secondary-side parallel resonant capacitor C2 and the rectifier diode D0.
As the level of withstand voltage of these components is lowered, for example, the components are reduced in size and the switching characteristics of switching components such as rectifier diodes are enhanced, thereby leading to a reduction of switching loss, for example. Therefore, if the withstand voltage of the components can be lowered for example by controlling the level of voltage generated on the secondary side, the switching characteristics of the components will be improved. This means an improvement in the characteristics of the power supply circuit.
Moreover, as shown in FIGS. 13C and 13F, in the circuits of FIGS. 9 and 10, high-frequency ringing occurs in the rectified current I0 on the secondary side when the rectifier diode D0 is turned on, thereby emitting noise. Therefore, in practice, as shown on the secondary side of the circuit of FIG. 10, the power supply circuit needs to be provided with a snubber circuit comprising a capacitor Csn and a resistance Rsn in parallel with the rectifier diode D0, for example. However, provision of the snubber circuit will result in an increase in power loss.
Also, it is known that while the circuits of FIGS. 15 and 16 provided with a series resonant circuit on the secondary side variably controls switching frequency according to variation in the secondary-side direct-current output voltage E0 due to load variation to effect constant-voltage control, an abnormal operation in which ZVS (Zero Voltage Switching) is not performed occurs under conditions of medium load.
Such abnormal operation occurs in the power supply circuit of FIG. 10 because the period TOFF during which the switching device Q1 is turned off is extended as the load power Po is decreased and the switching frequency is increased. During a period Ti during which such abnormal operation occurs, the switching device Q1 performs switching operation while having a certain level of voltage and current, and therefore power loss in the switching device Q1 is increased. Thus, it is necessary to enlarge a heat radiator for controlling heating of the switching device Q1.
In cases where the power supply circuit changes the switching frequency on the primary side and thus stabilizes the secondary-side direct-current output voltage by the complex control method, when an abnormal condition of a short circuit in the load on the secondary side occurs, the control system of the power supply circuit operates so as to lower the switching frequency. In a condition of a low switching frequency, the period TON during which the switching device is turned on is lengthened, and therefore the level of voltage (V1) and current (IQ1 and Icr) applied to the switching device Q1 and the parallel resonant capacitor Cr, for example, is increased.
Thus, in order to deal with a short-circuited load, it is necessary to provide the power supply circuit with an overvoltage protective circuit and an overcurrent protective circuit for protecting the switching device by limiting a high level of voltage and current generated at the time of the short circuit. Provision of the overvoltage protective circuit and the overcurrent protective circuit also hinders reduction of size and cost of the power supply circuit.
It is therefore an object of the present invention to provide a switching power supply circuit which is improved in its power conversion efficiency and is miniaturized.
In carrying out the invention and according to one aspect thereof, there is provided a switching power supply circuit comprising: an insulating converter transformer including a primary winding and a secondary winding insulated from each other, the primary winding and the secondary winding being loosely coupled to each other; a switching circuit including a switching device for performing switching operation on a current flowing into the primary winding of the insulating converter transformer at a fixed frequency; a primary-side parallel resonant capacitor provided on a primary side of the insulating converter transformer for forming a primary-side parallel resonant circuit in conjunction with inductance of the converter transformer; a secondary-side resonant capacitor provided on a secondary side of the insulating converter transformer for forming a secondary-side resonant circuit in conjunction with inductance of the converter transformer; a rectifier circuit for rectifying an alternating voltage obtained on the secondary side of the insulating converter transformer; an active clamp circuit provided on the secondary side of the insulating converter transformer for clamping the alternating voltage obtained on the secondary side of the insulating converter transformer in synchronism with the switching operation of the switching circuit; and a constant-voltage control circuit for effecting control for constant voltage by controlling a clamping period of the active clamp circuit and controlling an on/off-period duty ratio of a rectifying device of the rectifier circuit according to a level of output voltage of the rectifier circuit.
The configuration described above is that of a so-called complex resonance type switching converter provided with a primary-side parallel resonant circuit for forming a voltage resonance type converter on the primary side and a secondary-side resonant circuit formed by a secondary winding and a secondary-side resonant capacitor on the secondary side. The switching frequency of the voltage resonance type converter on the primary side is fixed at a given frequency.
On the basis of this configuration, an active clamp means for controlling a level of voltage obtained in the secondary winding is provided on the secondary side. Control for constant voltage is effected by allowing switching frequency to be dependent on the primary-side switching converter and therefore to be constant, and by variably controlling an on/off-period duty ratio of a secondary-side rectifier diode.
The above and other objects, features and advantages of the present invention will become apparent from the following description and the appended claims, taken in conjunction with the accompanying drawings in which like parts or elements denoted by like reference symbols.